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SLUU467–DECEMBER 2010
PR2000: A 90-W, High-Efficiency, LLC Series-Resonant Converter with
Secondary-Side Synchronous Rectification
Power Management – Consumer Isolated Power
1 INTRODUCTION
This guide documents the design of a low-profile, high-efficiency, LLC series-resonant DC/DC converter that
incorporates secondary-side synchronous rectifiers (SR). The converter is optimized for a 90-W laptop adapter
application and designed to operate from the high-voltage output produced by an upstream AC/DC boost power
factor correction (PFC) converter. The boost PFC converter would allow this adapter to operate from a universal
line-voltage input. The LLC resonant converter provides an isolated output of 19.5 VDC from an input voltage range
of 320–420 VDC. At a rating of 90 W the circuit has a maximum continuous load of 4.6 A.
Off-line ac adapters used for powering laptop PCs demand increasingly higher operating efficiencies in ever smaller
packages. The combination of high efficiency operation and a low-profile package reduces the adapter’s overall
size, weight and cost by minimizing the need for thermal management. The improved efficiency of this design is
made possible by replacing the Schottky rectifiers that are normally used in the secondary circuit with
synchronously controlled MOSFETs. Due to their low drain-source ‘on’ resistance, synchronously switched
MOSFETs can operate with a much lower voltage drop than regular diode rectifiers. Depending on the combination
of load current and output voltage the power dissipation of an adapter can be reduced by several watts using this
design.
The circuit features four integrated circuit devices from Texas Instruments. They include the UCC25600; a low-cost
resonant converter controller, and the UCC24610; a green rectifier controller. Other parts used include the TL431A
shunt regulator and the TPS71550 low drop-out linear regulator. The circuit requires a 12-VDC external bias supply
to operate. In a regular adapter design the bias power would be produced by the boost PFC AC/DC converter stage
that would normally precede this circuit.
2 SCOPE
The UCC24610 Green Rectifier Controller is optimized for 5-V systems and can be used for LLC outputs up to 15 V
when a separate 5-V supply is available. Above 15 V the UCC24610 is limited by the 50-V maximum voltage rating
of the VD pin. This is because in a conventional secondary rectifier arrangement, that employs two rectifiers with a
center-tapped secondary winding, each rectifier sees a peak reverse voltage equal to twice the regulated output.
The scope of this reference design guide is to describe the design and performance of a functional circuit that
extends the application of the UCC24610 to systems with output voltages up to 30 V. This is achieved using an
alternate topology for secondary rectification and addressing the design constraints that the topology presents. Two
configurations are described for synchronizing the turn-off of each SR circuit using the gate-drive signals on the
primary side of the converter.
An area not addressed by this guide is electromagnetic compatibility (EMC). For most applications, EMI filter
components are added so that the design meets applicable environmental and system compatibility requirements.
To comply with EMC standards, components such as input and output filters are required to suppress
electromagnetic interference (EMI).
SLUU467–DECEMBER 2010
3 ELECTRICAL PERFORMANCE
Table 1 Performance Specifications
Symbol Parameter Notes and Conditions Min Nom Max Units
INPUT CHARACTERSTICS
VI Input Voltage 320 420 VDC
II Input Current 0.5 A
VCC Bias supply voltage 11.5 16 V
ICC Bias supply current Output enabled 20 mA
OUTPUT CHARACTERSTICS
VO Output Voltage 19.2 19.5 19.
(1)
V
IO Output Current 1
(2)
4.6 A
PO Output Power 90 W
ILIM Current Limit ΔVO = -2 V 6 A
ΔVLOAD Load Regulation VI =390 V 0.05 %V O
ΔVLINE Line Regulation^ IO =3 A^ 0.
VO(ripple) Output Voltage Ripple V I =390 V, I O =3 A 150 mVPP
SYSTEM CHARACTERSTICS
η Efficiency VI =390 V, I O =3 A 94 %
Overall thickness
(3)
18 mm
Temp. Range Nat’l Conv. airflow 0 50 °C
(1) Equivalent to an output voltage tolerance of ±1.5%.
(2) Operates at no load with reduced regulation and burst mode operation.
(3) Excludes terminal blocks for power input and output connections.
SLUU467–DECEMBER 2010
4.2 Secondary Rectifier Configuration
LLC converters generally use a center-tapped transformer secondary winding with two rectifiers; one at each end of
the winding. This allows the rectifiers and each half winding to share the load current by conducting alternate half
cycles. However the rectifiers must be rated at more than 2× the output voltage.
To limit the peak voltage to the VD pin of the UCC24610 SR controller, the rectifiers in this design are stacked in a
voltage doubling arrangement. This reduces the voltage seen by each rectifier to 1× the output voltage, allowing
MOSFETS with a lower VDS (max) and on resistance to be used. It also eliminates the requirement for a center tap
on the main transformer secondary winding, reducing manufacturing cost. However there are compromises. Both
rectifiers are now required to conduct the full load current. This doubles the peak current in the secondary winding,
which must now pass current in both half cycles. The combination of twice the current magnitude and full-cycle
conduction increases the rms current in the secondary winding by a factor of 2 × √2. Secondly the rectifiers no
longer share a common anode, requiring the control circuit for the upper rectifier (D1 in Figure 2) to float on the ac-
voltage end of the transformer secondary winding. This is the switch node between the two rectifiers.
Figure 2 Voltage-Doubling Rectifier Circuit with Synchronously Switched MOSFETs
SLUU467–DECEMBER 2010
5 SCHEMATICS AND CIRCUIT DESCRIPTION
The schematics provided in this section are for reference only. For the purposes of clarity some of the detailed
component parameters are not shown. Consult the list of materials for additional information.
Three versions of the schematic are presented. The choice depends on the desired synchronization method for
turning the UCC24610 SR controllers off, using the gate-drive signals from the primary side of the converter. The
synchronization configurations are described in detail in the section 5.2, Secondary-Side Synchronous Rectifier
Circuit. The primary circuit and feedback control is the same for all three synchronization methods. The following
description refers to Figure 3, Figure 4, and Figure 5.
5.1 Primary Circuit and Feedback Control
The half-bridge power stage is comprised of the MOSFETs Q1, Q2. These are controlled by the UCC25600 via an
isolated gate drive, T2. The UCC25600 is a very simple and low-cost part to use. In addition to providing a variable
drive frequency for the ½-bridge power stage, it includes a soft-start feature, over-current shutdown protection, and
adjustment of the switch dead time between Q1 and Q2.
Load current is sensed through the resonant capacitance using a parallel 0.001-μF capacitor to ground (C7). This
capacitor is in parallel with the 0.02-μF of resonant capacitance (C5+C6) to form a 1:21 impedance divider. Current
through C7 is half-wave rectified (D3) and then passed through a sense resistor (R19). The resulting signal is then
filtered (R18 and C25) and fed to the OC pin (pin 3) of the UCC25600 (U1). This method of sensing current
provides less variation in the sensed current versus the switching frequency.
The resonant current is susceptible to a high surge current during converter start-up. For this reason it is
recommended that the sensitivity of the current sense circuit is reduced during converter start up. This circuit
accomplishes this using a p-channel JFET (Q6) and a divider resistor (R20). The gate of the JFET is connected to
the SS pin (pin 4) of the UCC25600, which is initially low during converter start up. The JFET places the divider
resistor in circuit during the period that its gate voltage is low. When the soft-start period is complete the rising gate
voltage to the JFET pinches off its channel. This isolates the divider resistor and returns the current sense gain to
normal.
A limitation of LLC series-resonant converters is that they operate over a limited input voltage range. This is
because below a certain operating frequency the frequency-gain relationship of the converter is reversed. The
operating frequency of the UCC25600 is controlled by the magnitude of current flowing from the RT pin (pin 2). A
resistor to ground, R22, sets the minimum operating frequency to approximately 70 kHz.
To help limit the surge current during start-up the UCC25600 incorporates a soft-start feature. The maximum
control frequency is set by R23. The value of this resistor sets the maximum frequency of the UCC25600 above
360 kHz. Above 360 kHz the device will enter burst-mode in order to maintain control of the output at light loads.
The converter output voltage is regulated by a TL431A shunt regulator (U7) located in the secondary. The error
signal generated from the TL431A is passed back to the converter primary using a coupler (U6). A decrease in the
voltage at the output of the coupler increases the current pulled from RC pin (pin 2) of the UCC25600. This
increases the switching frequency of the converter to reduce its gain and output voltage.
5.2 Secondary-Side Synchronous Rectifier Circuit
The control circuit for the two synchronous rectifiers (SRs) in the secondary is identical. The only difference is the
method by which the circuits are powered. The upper SR circuit formed by U2, U4, Q3 and C19 effectively sits on
top of the lower SR circuit formed by U3, U5, Q4, and C20. The voltage developed across each circuit’s capacitor
(C19 and C20) is approximately 9.75 VDC, producing a total of 19.5 VDC across both capacitors. The supply
voltage to each UCC24610 controller (U4, U5) is tightly regulated to 5 VDC by a low-cost linear regulator (U2, U3).
The lower SR circuit is referenced to secondary ground (SECGND) and can be powered directly from the steady-
state 9.75 VDC produced across C20. However the upper SR circuit floats on the 10 VAC produced at one end of
the transformer secondary winding. The voltage at this node swings approximately ±10 V with respect to the
junction of C19 and C20, requiring the upper circuit to be powered using the bootstrap diode (D1). During operation
SLUU467–DECEMBER 2010
Figure 3 Schematic, UCC24610 LLC Series-Resonant Converter – Self Synchronization Configuration
SLUU467–DECEMBER 2010
Figure 4 Schematic, UCC24610 LLC Series-Resonant Converter – Transformer Configuration
SLUU467–DECEMBER 2010
6 LIST OF MATERIALS
The component list for the three build configurations described herein is defined by combining the contents of up to
two of the three tables in this section. Table 2 lists the components for the “self-synchronization” configuration. The
self-synchronization configuration has the minimum component count and is common to all three configurations.
For this reason it is defined as the base circuit for the “Transformer Synchronization” and “Capacitor
Synchronization” configurations. The following summary identifies the material list for each configuration.
1. Self-Synchronization: Use only Table 2
2. Transformer Synchronization: Combine Table 2 with the changes outlined in Table 3.
3. Capacitor Synchronization: Combine Table 2 with the changes outlined in Table 4.
Table 2 Components for Self-Synchronization Configuration (Base Circuit)
Ref. Value Tol. Description Part Number Manufacturer
C1 1.0 μF 10% Capacitor, Film (PP), 630 V B32674D6105K EPCOS
C2 N/A Not Fitted
C4 0.1 μF 10% Capacitor, MLC, 0805, 25 V Generic Multi-Sourced
C5 0.01 μF 5% Capacitor, Film (PP), 630 V B32621A6103J EPCOS C6 0.01 μF 5% Capacitor, Film (PP), 630 V B32621A6103J EPCOS
C7 0.001 μF 10% Capacitor, Film, (PP), 1600 V MKP10 1000/1600/10 WIMA
C8 N/A Not Fitted
C9 0.33 μF 20% Capacitor, MLC, 0805, 50 V Generic Multi-Sourced C10 0.33 μF 20% Capacitor, MLC, 0805, 50 V Generic Multi-Sourced
C11 1 μF 10% Capacitor, MLC, 0805, 16 V Generic Multi-Sourced
C12 1 μF 10% Capacitor, MLC, 0805, 16 V Generic Multi-Sourced
C13 N/A Not Fitted C14 N/A Not Fitted
C15 N/A Not Fitted
C16 N/A Not Fitted
C17 47 pF 5% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced C18 47 pF 5% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced
C19 470 μF 20% Capacitor, Electrolytic, 16 V ORZ471M1CSA-10127S Surge Components Inc.
C20 470 μF 20% Capacitor, Electrolytic, 16 V ORZ471M1CSA-10127S Surge Components Inc.
C21 4.7 μF 20% Capacitor, MLC, 1210, 50 V Generic Multi-Sourced C22 0.1 μF 20% Capacitor, MLC, 1206, 50 V Generic Multi-Sourced
C23 0.22 μF 10% Capacitor, MLC, 0603, 10 V Generic Multi-Sourced
C24 1 μF 10% Capacitor, MLC, 0805, 16 V Generic Multi-Sourced
C25 0.01 μF 10% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced C26 0.047 μF 10% Capacitor, MLC, 0603, 25 V Generic Multi-Sourced
C27 N/A Not Fitted
C28 N/A Not Fitted
C29 0.33 μF 5% Capacitor, MLC, 0805, 50 V Generic Multi-Sourced
SLUU467–DECEMBER 2010
Ref. Value Tol. Description Part Number Manufacturer
D1 Diode, Switching, 100 V, 200 mA MMBD914 Multi-Sourced
D3 Diode, Switching, 100 V, 200 mA MMBD914 Multi-Sourced
D4 Diode, Switching, 100 V, 200 mA MMBD914 Multi-Sourced
D5 Diode, Schottky, 70 V, 15 mA 1N5711 ST Microelectronics
D6 Diode, Zener, 0.5-W, 12 V MMSZ5242BT1 ON Semiconductor
J1 Term Block, Eurostyle, 3-Way 39390-0103 Molex
J2 Terminal Block, 2-Way ED120/2DS On-Shore Technology
J3 Header, 0.025 Sq., 2-Way 68001-203HLF FCI BergStik
J4 Terminal Block, 4-Way ED120/4DS On Shore Technology
Q1 N-MOSFET, TO-220, 600 V, 7 A STF9NK60ZD ST Microelectronics
Q2 N-MOSFET, TO-220, 600 V, 7 A STF9NK60ZD ST Microelectronics
Q3 N-MOSFET, LF-PACK, 60 V, 50 A BSC067N06LS3 G Infineon
Q4 N-MOSFET, LF-PACK, 60 V, 50 A BSC067N06LS3 G Infineon
Q5 TRANSISTOR, NPN, 40 V, 200 mA MMBT3904 Multi-Sourced
Q6 JFET, P-CHL, 30 V, 50 mA MMBFJ176 Fairchild
R1 15R0 1% Resistor, SMD, 0805 Generic Multi-Sourced
R2 15R0 1% Resistor, SMD, 0805 Generic Multi-Sourced
R3 N/A Not Fitted
R4 N/A Not Fitted
R5 49k9 1% Resistor, SMD, 0603 Generic Multi-Sourced
R6 49k9 1% Resistor, SMD, 0603 Generic Multi-Sourced
R7 274k 1% Resistor, SMD, 0603 RK73H1JTTD2743F KOA
R8 274k 1% Resistor, SMD, 0603 RK73H1JTTD2743F KOA
R9 49R9 1% Resistor, SMD, 0805 RK73H2ATTD49R9F KOA
R10 49R9 1% Resistor, SMD, 0805 RK73H2ATTD49R9F KOA
R11 49R9 1% Resistor, SMD, 0603 Generic Multi-Sourced
R12 49R9 1% Resistor, SMD, 0603 Generic Multi-Sourced
R13 2R49 1% Resistor, SMD, 0603 Generic Multi-Sourced
R14 2R49 1% Resistor, SMD, 0603 Generic Multi-Sourced
R15 31k6 1% Resistor, SMD, 0603 Generic Multi-Sourced
R16 60k4 1% Resistor, SMD, 0603 Generic Multi-Sourced
R17 10k 1% Resistor, SMD, 0603 Generic Multi-Sourced
R18 10k 1% Resistor, SMD, 0603 Generic Multi-Sourced
R19 49R9 1% Resistor, SMD, 0805 RK73H2ATTD49R9F KOA
R20 10k 1% Resistor, SMD, 0603 Generic Multi-Sourced
R21 26k1 1% Resistor, SMD, 0603 RK73H1JTTD2612F
R22 3k65 1% Resistor, SMD, 0603 Generic Multi-Sourced
R23 348R 1% Resistor, SMD, 0603 Generic Multi-Sourced
R24 6k49 1% Resistor, SMD, 0603 Generic Multi-Sourced
SLUU467–DECEMBER 2010
Table 3 Component Changes for Transformer Synchronization Configuration
Ref. Value Tol. Description Part Number Manufacturer C2 N/A Not Fitted
C8 0.1 μF 10% Capacitor, MLC, 0805, 25 V Generic Multi-Sourced
C13 1000 pF 10% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced
C14 1000 pF 10% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced C15 N/A Not Fitted
C16 N/A Not Fitted
R3 10R0 1% Resistor, SMD, 0603 Generic Multi-Sourced
R4 10R0 1% Resistor, SMD, 0603 Generic Multi-Sourced R31 0R0 Link, SMD, 0805 Generic Multi-Sourced
R32 1k 1% Resistor, SMD, 0603 Generic Multi-Sourced
R33 1k 1% Resistor, SMD, 0603 Generic Multi-Sourced
T3 Pulse Transformer (2:1:1), SMD PA2008NL Pulse
Table 4 Component Changes for Capacitor Synchronization Configuration
Ref. Value Tol. Description Part Number Manufacturer C2 1,000 pF 20% Capacitor, Ceramic T/H, 500 VAC VY1102M35Y5UQ63V0 Vishay-BCcomponents
C8 0R0 Link, SMD, 0805 Generic Multi-Sourced
C13 0R0 Link, SMD, 0603 Generic Multi-Sourced
C14 0R0 Link, SMD, 0603 Generic Multi-Sourced C15 47 pF 10% Capacitor, Ceramic T/H, 500 VAC VY1470K31Y5SQ63V0 Vishay-BCcomponents
C16 47 pF 10% Capacitor, Ceramic T/H, 500 VAC VY1470K31Y5SQ63V0 Vishay-BCcomponents
R3 475R 1% Resistor, SMD, 0603 Generic Multi-Sourced
R4 475R 1% Resistor, SMD, 0603 Generic Multi-Sourced R31 0R0 Link, SMD, 0805 Generic Multi-Sourced
R32 N/A Not Fitted
R33 N/A Not Fitted
T3 N/A Not Fitted
SLUU467–DECEMBER 2010
7 PCB DESIGN
7.1 General
Figure 6 and Figure 7 show the component placement and copper routing of the printed circuit board (PCB) used to
test and characterize this design. The PCB design is based on a double-sided 2-oz. copper-foil layout. A double-
sided PCB provides the flexibility to optimize the layout of the SR FETs and control circuitry. A combination of both
through-hole (T/H) and surface mount devices (SMD) were placed on the top side, and SMD parts on the foil side.
7.2 Grounding
High-frequency transient current associated with switched-power converters induce differential noise voltages in the
ground system. Any voltage differential in the ground can cause spurious operation of the control circuits. The
design dedicates large areas of copper to power ground (PWRGND) and analog ground (AGND). PWRGND
provides the return path for the power circuitry on the primary side of the converter. AGND is used by the control
circuits as the primary-side zero-volt reference. To prevent the AGND from being used as current path by high-
frequency power signals, AGND is referenced to PWRGND at just one location. This single-point grounding
technique forces the high-frequency ground currents generated by the power circuitry to be directed around (as
opposed to through) the quiet ground area of the sensitive control circuits.
It is important to keep the ground connections contiguous. Ground connections should take priority over the routing
of other signals. Where necessary, use vias and T/H components to pass signals to other areas of the board.
These techniques minimize the impedance to high-frequency ground currents to ensure low-noise and reliable
operation.
7.3 Creepage and Clearance
Both the component placement and spacing between the copper areas or the PCB were designed to comply with
the creepage and clearance requirements defined in the UL 60950 safety standard. The board was designed to
meet the requirements for functional isolation for the high-voltage nodes on the primary (ac-line) side of the power
supply, and reinforced isolation between all primary and secondary circuits. To comply with the UL standard 4 mm
of separation was used for the primary-side high-voltage traces, and 8 mm between all primary and secondary side
traces. The default trace separation for low-voltage nodes, as used for control circuits, was 0.3 mm.
7.4 Thermal Considerations
Although this circuit and topology operates with high efficiency, some components on the PCB assembly dissipate
a moderate amount of heat. This may require thermal management initiatives to be implemented when the
converter operates in an enclosure. A thermally conductive path to the surfaces of the enclosure may be required
for these components. Table 5 provides a list of the components that have significant heat dissipation.
Table 5 Parts with High Thermal Dissipation
Reference
Designator Description Dissipation
(1)
Q3, Q4 Secondary SR FETs 1 W
T1 LLC converter transformer 3 W
(1) Estimated dissipation at maximum load and 390 V input voltage.
SLUU467–DECEMBER 2010
Figure 7 PCB Layout; Foil-Side View
SLUU467–DECEMBER 2010
7.6 Power Inputs
7.6.1 HVDC Input
The power input to the circuit requires a high-voltage DC source, capable of supplying up to 420 VDC at 0.5 A max.
This can be provided by a high-voltage adjustable bench supply, the output from a boost PFC circuit, or the rectified
output from a 0–280 V, 1-kVA rated power rheostat. Ensure that the source is either current limited or is fitted with a
fuse with a rating of no more than 1 A.
The HVDC power can be safely applied without the bias supply being present.
7.6.2 Bias Supply
A bias supply of 12–16 VDC at 50 mA is required to operate the converter. The bias power is applied to the J
terminal block and can be supplied from a small adjustable bench supply or 12-V lamp battery. The current
consumption is typically 20 mA when the converter is operating. The polarity is correct when the negative
connection is closest to the J3 connector.
7.7 J3 Header
The PCB design includes a 0.1-in spaced pin header, J3. This header provides on/off control of the converter when
both HVDC input and bias power is applied. This interface can be used to hold down the SS pin (pin 4) until there is
sufficient output voltage for the converter to produce a regulated output. The converter runs only when the open-
circuit voltage at pin 1 of J3 is pulled to GND (J3/Pin 2) using either a jumper or NPN bipolar transistor.
Place a standard 2-pin shorting jumper at J3 to enable the converter to run and produce a DC output. When the
converter is enabled the UCC25600 resonant controller initiates a soft start. Removal of the J3 jumper promptly
shuts down the converter.
The application and removal of the bias supply has the same effect as the removal and replacement of the shorting
jumper at J3.
7.8 Monitoring Primary Current
By making a small modification to the PCB the primary current can be monitored using a current probe. This is also
the current flowing between the series-resonant components, L R and C R.
The PCB layout in Figure 6 shows two test points labeled “SWNODE” located close to and just above the primary
switch, Q2. They are identified TP3 and TP4 on the schematic. The 0.05-in trace connecting these two test points
carries the transformer primary current. This trace can easily be cut, and replaced with a loop of wire connecting
TP3 and TP4. The primary current can then be measured using a clip-on current probe.
SLUU467–DECEMBER 2010
8.3 Typical Output AC Ripple Voltage
0.
0.
0.
0.
0.
0.
1.0 1.5 2.0 2.5 3.0 3.5 4.0 4. Output Current (A)
Output Ripple (V
PP )^330 360 390 420
V (^) IN (V)
Figure 12 Output AC-Ripple with UCC Rectifier Controllers and FETs (2)
0.
0.
0.
0.
0.
0.
1.0 1.5 2.0 2.5 3.0 3.5 4.0 4. Output Current (A)
Output Ripple (V
PP )
330 360 390 420
V (^) IN (V)
Figure 13 Output AC-Ripple with MBR Schottky Rectifiers (2),^ (3)
(2) Peak-to-peak output ripple derived from V (^) OUT rms measurement using an oscilloscope with 20-MHz bandwidth limit. (3) The Schottky diode rectifiers show a higher magnitude of ac rippled at high input voltage (420 V). This is due to an increase in high- frequency ringing from these devices at this input voltage.
9 CIRCUIT WAVEFORMS
The oscilloscope waveforms in this section, captured in Figure 14 through Figure 29, can be identified using
the same color for source identification. The color key to the signals is provided in Table 6.
Table 6 Color Key for Oscilloscope Waveforms
Trace
Color Signal Description
Black Main transformer secondary winding output voltage
Blue Main transformer primary current
Green Upper SR GATE signal waveform (U4/pin 5)
Brown Upper SR SYNC signal waveform (U4/pin 1)
Red Lower SR GATE signal waveform (U5/pin 5)
Orange Lower SR SYNC signal waveform (U5/pin 1)
SLUU467–DECEMBER 2010
9.1 SR Synchronization with Pulse Transformer
Figure 14 Upper and Lower SR Gate Signals at 320-VDC and 1-A Load (f (^) SW <f (^) R )
Figure 15 Upper and Lower Gate Signals at 400-VDC Input and 4.6-A Load (f (^) SW ≈ fR )
Figure 16 Upper and Lower SR Gate Signals at 420-VDC Input and 4.6-A Load (f (^) SW >fR )
Figure 17 Upper and Lower SR Gate Signals at 420-VDC Input and 1-A Load (fSW >fR )
Figure 18 Upper/Lower SR Gate and Lower SYNC Signals at 340-VDC Input and 1-A Load (1)/
Figure 19 Upper SR Gate and SYNC Signals at 420-VDC and 3-A Load (2)/
(1) Below resonance the lower SR controller has set the gate signal (Red) low before the negative SYNC pulse (Orange). (2) Above resonance the upper SR controller sets the gate signal (Green) low about the same time as the SYNC pulse (Brown). This forces the gate output low irrespective of whether the UCC24610 controller has sensed the FET is no longer conducting.
