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A 90 w high efficiency, lcc series resonant converter, Study Guides, Projects, Research of Power Electronics

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Reference Design
SLUU467–DECEMBER 2010
PR2000: A 90-W, High-Efficiency, LLC Series-Resonant Converter with
Secondary-Side Synchronous Rectification
Power Management – Consumer Isolated Power
1 INTRODUCTION
This guide documents the design of a low-profile, high-efficiency, LLC series-resonant DC/DC converter that
incorporates secondary-side synchronous rectifiers (SR). The converter is optimized for a 90-W laptop adapter
application and designed to operate from the high-voltage output produced by an upstream AC/DC boost power
factor correction (PFC) converter. The boost PFC converter would allow this adapter to operate from a universal
line-voltage input. The LLC resonant converter provides an isolated output of 19.5 VDC from an input voltage range
of 320–420 VDC. At a rating of 90 W the circuit has a maximum continuous load of 4.6 A.
Off-line ac adapters used for powering laptop PCs demand increasingly higher operating efficiencies in ever smaller
packages. The combination of high efficiency operation and a low-profile package reduces the adapter’s overall
size, weight and cost by minimizing the need for thermal management. The improved efficiency of this design is
made possible by replacing the Schottky rectifiers that are normally used in the secondary circuit with
synchronously controlled MOSFETs. Due to their low drain-source ‘on’ resistance, synchronously switched
MOSFETs can operate with a much lower voltage drop than regular diode rectifiers. Depending on the combination
of load current and output voltage the power dissipation of an adapter can be reduced by several watts using this
design.
The circuit features four integrated circuit devices from Texas Instruments. They include the UCC25600; a low-cost
resonant converter controller, and the UCC24610; a green rectifier controller. Other parts used include the TL431A
shunt regulator and the TPS71550 low drop-out linear regulator. The circuit requires a 12-VDC external bias supply
to operate. In a regular adapter design the bias power would be produced by the boost PFC AC/DC converter stage
that would normally precede this circuit.
2 SCOPE
The UCC24610 Green Rectifier Controller is optimized for 5-V systems and can be used for LLC outputs up to 15 V
when a separate 5-V supply is available. Above 15 V the UCC24610 is limited by the 50-V maximum voltage rating
of the VD pin. This is because in a conventional secondary rectifier arrangement, that employs two rectifiers with a
center-tapped secondary winding, each rectifier sees a peak reverse voltage equal to twice the regulated output.
The scope of this reference design guide is to describe the design and performance of a functional circuit that
extends the application of the UCC24610 to systems with output voltages up to 30 V. This is achieved using an
alternate topology for secondary rectification and addressing the design constraints that the topology presents. Two
configurations are described for synchronizing the turn-off of each SR circuit using the gate-drive signals on the
primary side of the converter.
An area not addressed by this guide is electromagnetic compatibility (EMC). For most applications, EMI filter
components are added so that the design meets applicable environmental and system compatibility requirements.
To comply with EMC standards, components such as input and output filters are required to suppress
electromagnetic interference (EMI).
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pf5
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Download A 90 w high efficiency, lcc series resonant converter and more Study Guides, Projects, Research Power Electronics in PDF only on Docsity!

SLUU467–DECEMBER 2010

PR2000: A 90-W, High-Efficiency, LLC Series-Resonant Converter with

Secondary-Side Synchronous Rectification

Power Management – Consumer Isolated Power

1 INTRODUCTION

This guide documents the design of a low-profile, high-efficiency, LLC series-resonant DC/DC converter that

incorporates secondary-side synchronous rectifiers (SR). The converter is optimized for a 90-W laptop adapter

application and designed to operate from the high-voltage output produced by an upstream AC/DC boost power

factor correction (PFC) converter. The boost PFC converter would allow this adapter to operate from a universal

line-voltage input. The LLC resonant converter provides an isolated output of 19.5 VDC from an input voltage range

of 320–420 VDC. At a rating of 90 W the circuit has a maximum continuous load of 4.6 A.

Off-line ac adapters used for powering laptop PCs demand increasingly higher operating efficiencies in ever smaller

packages. The combination of high efficiency operation and a low-profile package reduces the adapter’s overall

size, weight and cost by minimizing the need for thermal management. The improved efficiency of this design is

made possible by replacing the Schottky rectifiers that are normally used in the secondary circuit with

synchronously controlled MOSFETs. Due to their low drain-source ‘on’ resistance, synchronously switched

MOSFETs can operate with a much lower voltage drop than regular diode rectifiers. Depending on the combination

of load current and output voltage the power dissipation of an adapter can be reduced by several watts using this

design.

The circuit features four integrated circuit devices from Texas Instruments. They include the UCC25600; a low-cost

resonant converter controller, and the UCC24610; a green rectifier controller. Other parts used include the TL431A

shunt regulator and the TPS71550 low drop-out linear regulator. The circuit requires a 12-VDC external bias supply

to operate. In a regular adapter design the bias power would be produced by the boost PFC AC/DC converter stage

that would normally precede this circuit.

2 SCOPE

The UCC24610 Green Rectifier Controller is optimized for 5-V systems and can be used for LLC outputs up to 15 V

when a separate 5-V supply is available. Above 15 V the UCC24610 is limited by the 50-V maximum voltage rating

of the VD pin. This is because in a conventional secondary rectifier arrangement, that employs two rectifiers with a

center-tapped secondary winding, each rectifier sees a peak reverse voltage equal to twice the regulated output.

The scope of this reference design guide is to describe the design and performance of a functional circuit that

extends the application of the UCC24610 to systems with output voltages up to 30 V. This is achieved using an

alternate topology for secondary rectification and addressing the design constraints that the topology presents. Two

configurations are described for synchronizing the turn-off of each SR circuit using the gate-drive signals on the

primary side of the converter.

An area not addressed by this guide is electromagnetic compatibility (EMC). For most applications, EMI filter

components are added so that the design meets applicable environmental and system compatibility requirements.

To comply with EMC standards, components such as input and output filters are required to suppress

electromagnetic interference (EMI).

SLUU467–DECEMBER 2010

3 ELECTRICAL PERFORMANCE

Table 1 Performance Specifications

Symbol Parameter Notes and Conditions Min Nom Max Units

INPUT CHARACTERSTICS

VI Input Voltage 320 420 VDC

II Input Current 0.5 A

VCC Bias supply voltage 11.5 16 V

ICC Bias supply current Output enabled 20 mA

OUTPUT CHARACTERSTICS

VO Output Voltage 19.2 19.5 19.

(1)

V

IO Output Current 1

(2)

4.6 A

PO Output Power 90 W

ILIM Current Limit ΔVO = -2 V 6 A

ΔVLOAD Load Regulation VI =390 V 0.05 %V O

ΔVLINE Line Regulation^ IO =3 A^ 0.

VO(ripple) Output Voltage Ripple V I =390 V, I O =3 A 150 mVPP

SYSTEM CHARACTERSTICS

η Efficiency VI =390 V, I O =3 A 94 %

Overall thickness

(3)

18 mm

Temp. Range Nat’l Conv. airflow 0 50 °C

(1) Equivalent to an output voltage tolerance of ±1.5%.

(2) Operates at no load with reduced regulation and burst mode operation.

(3) Excludes terminal blocks for power input and output connections.

SLUU467–DECEMBER 2010

4.2 Secondary Rectifier Configuration

LLC converters generally use a center-tapped transformer secondary winding with two rectifiers; one at each end of

the winding. This allows the rectifiers and each half winding to share the load current by conducting alternate half

cycles. However the rectifiers must be rated at more than 2× the output voltage.

To limit the peak voltage to the VD pin of the UCC24610 SR controller, the rectifiers in this design are stacked in a

voltage doubling arrangement. This reduces the voltage seen by each rectifier to 1× the output voltage, allowing

MOSFETS with a lower VDS (max) and on resistance to be used. It also eliminates the requirement for a center tap

on the main transformer secondary winding, reducing manufacturing cost. However there are compromises. Both

rectifiers are now required to conduct the full load current. This doubles the peak current in the secondary winding,

which must now pass current in both half cycles. The combination of twice the current magnitude and full-cycle

conduction increases the rms current in the secondary winding by a factor of 2 × √2. Secondly the rectifiers no

longer share a common anode, requiring the control circuit for the upper rectifier (D1 in Figure 2) to float on the ac-

voltage end of the transformer secondary winding. This is the switch node between the two rectifiers.

Figure 2 Voltage-Doubling Rectifier Circuit with Synchronously Switched MOSFETs

SLUU467–DECEMBER 2010

5 SCHEMATICS AND CIRCUIT DESCRIPTION

The schematics provided in this section are for reference only. For the purposes of clarity some of the detailed

component parameters are not shown. Consult the list of materials for additional information.

Three versions of the schematic are presented. The choice depends on the desired synchronization method for

turning the UCC24610 SR controllers off, using the gate-drive signals from the primary side of the converter. The

synchronization configurations are described in detail in the section 5.2, Secondary-Side Synchronous Rectifier

Circuit. The primary circuit and feedback control is the same for all three synchronization methods. The following

description refers to Figure 3, Figure 4, and Figure 5.

5.1 Primary Circuit and Feedback Control

The half-bridge power stage is comprised of the MOSFETs Q1, Q2. These are controlled by the UCC25600 via an

isolated gate drive, T2. The UCC25600 is a very simple and low-cost part to use. In addition to providing a variable

drive frequency for the ½-bridge power stage, it includes a soft-start feature, over-current shutdown protection, and

adjustment of the switch dead time between Q1 and Q2.

Load current is sensed through the resonant capacitance using a parallel 0.001-μF capacitor to ground (C7). This

capacitor is in parallel with the 0.02-μF of resonant capacitance (C5+C6) to form a 1:21 impedance divider. Current

through C7 is half-wave rectified (D3) and then passed through a sense resistor (R19). The resulting signal is then

filtered (R18 and C25) and fed to the OC pin (pin 3) of the UCC25600 (U1). This method of sensing current

provides less variation in the sensed current versus the switching frequency.

The resonant current is susceptible to a high surge current during converter start-up. For this reason it is

recommended that the sensitivity of the current sense circuit is reduced during converter start up. This circuit

accomplishes this using a p-channel JFET (Q6) and a divider resistor (R20). The gate of the JFET is connected to

the SS pin (pin 4) of the UCC25600, which is initially low during converter start up. The JFET places the divider

resistor in circuit during the period that its gate voltage is low. When the soft-start period is complete the rising gate

voltage to the JFET pinches off its channel. This isolates the divider resistor and returns the current sense gain to

normal.

A limitation of LLC series-resonant converters is that they operate over a limited input voltage range. This is

because below a certain operating frequency the frequency-gain relationship of the converter is reversed. The

operating frequency of the UCC25600 is controlled by the magnitude of current flowing from the RT pin (pin 2). A

resistor to ground, R22, sets the minimum operating frequency to approximately 70 kHz.

To help limit the surge current during start-up the UCC25600 incorporates a soft-start feature. The maximum

control frequency is set by R23. The value of this resistor sets the maximum frequency of the UCC25600 above

360 kHz. Above 360 kHz the device will enter burst-mode in order to maintain control of the output at light loads.

The converter output voltage is regulated by a TL431A shunt regulator (U7) located in the secondary. The error

signal generated from the TL431A is passed back to the converter primary using a coupler (U6). A decrease in the

voltage at the output of the coupler increases the current pulled from RC pin (pin 2) of the UCC25600. This

increases the switching frequency of the converter to reduce its gain and output voltage.

5.2 Secondary-Side Synchronous Rectifier Circuit

The control circuit for the two synchronous rectifiers (SRs) in the secondary is identical. The only difference is the

method by which the circuits are powered. The upper SR circuit formed by U2, U4, Q3 and C19 effectively sits on

top of the lower SR circuit formed by U3, U5, Q4, and C20. The voltage developed across each circuit’s capacitor

(C19 and C20) is approximately 9.75 VDC, producing a total of 19.5 VDC across both capacitors. The supply

voltage to each UCC24610 controller (U4, U5) is tightly regulated to 5 VDC by a low-cost linear regulator (U2, U3).

The lower SR circuit is referenced to secondary ground (SECGND) and can be powered directly from the steady-

state 9.75 VDC produced across C20. However the upper SR circuit floats on the 10 VAC produced at one end of

the transformer secondary winding. The voltage at this node swings approximately ±10 V with respect to the

junction of C19 and C20, requiring the upper circuit to be powered using the bootstrap diode (D1). During operation

SLUU467–DECEMBER 2010

Figure 3 Schematic, UCC24610 LLC Series-Resonant Converter – Self Synchronization Configuration

SLUU467–DECEMBER 2010

Figure 4 Schematic, UCC24610 LLC Series-Resonant Converter – Transformer Configuration

SLUU467–DECEMBER 2010

6 LIST OF MATERIALS

The component list for the three build configurations described herein is defined by combining the contents of up to

two of the three tables in this section. Table 2 lists the components for the “self-synchronization” configuration. The

self-synchronization configuration has the minimum component count and is common to all three configurations.

For this reason it is defined as the base circuit for the “Transformer Synchronization” and “Capacitor

Synchronization” configurations. The following summary identifies the material list for each configuration.

1. Self-Synchronization: Use only Table 2

2. Transformer Synchronization: Combine Table 2 with the changes outlined in Table 3.

3. Capacitor Synchronization: Combine Table 2 with the changes outlined in Table 4.

Table 2 Components for Self-Synchronization Configuration (Base Circuit)

Ref. Value Tol. Description Part Number Manufacturer

C1 1.0 μF 10% Capacitor, Film (PP), 630 V B32674D6105K EPCOS

C2 N/A Not Fitted

C4 0.1 μF 10% Capacitor, MLC, 0805, 25 V Generic Multi-Sourced

C5 0.01 μF 5% Capacitor, Film (PP), 630 V B32621A6103J EPCOS C6 0.01 μF 5% Capacitor, Film (PP), 630 V B32621A6103J EPCOS

C7 0.001 μF 10% Capacitor, Film, (PP), 1600 V MKP10 1000/1600/10 WIMA

C8 N/A Not Fitted

C9 0.33 μF 20% Capacitor, MLC, 0805, 50 V Generic Multi-Sourced C10 0.33 μF 20% Capacitor, MLC, 0805, 50 V Generic Multi-Sourced

C11 1 μF 10% Capacitor, MLC, 0805, 16 V Generic Multi-Sourced

C12 1 μF 10% Capacitor, MLC, 0805, 16 V Generic Multi-Sourced

C13 N/A Not Fitted C14 N/A Not Fitted

C15 N/A Not Fitted

C16 N/A Not Fitted

C17 47 pF 5% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced C18 47 pF 5% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced

C19 470 μF 20% Capacitor, Electrolytic, 16 V ORZ471M1CSA-10127S Surge Components Inc.

C20 470 μF 20% Capacitor, Electrolytic, 16 V ORZ471M1CSA-10127S Surge Components Inc.

C21 4.7 μF 20% Capacitor, MLC, 1210, 50 V Generic Multi-Sourced C22 0.1 μF 20% Capacitor, MLC, 1206, 50 V Generic Multi-Sourced

C23 0.22 μF 10% Capacitor, MLC, 0603, 10 V Generic Multi-Sourced

C24 1 μF 10% Capacitor, MLC, 0805, 16 V Generic Multi-Sourced

C25 0.01 μF 10% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced C26 0.047 μF 10% Capacitor, MLC, 0603, 25 V Generic Multi-Sourced

C27 N/A Not Fitted

C28 N/A Not Fitted

C29 0.33 μF 5% Capacitor, MLC, 0805, 50 V Generic Multi-Sourced

SLUU467–DECEMBER 2010

Ref. Value Tol. Description Part Number Manufacturer

D1 Diode, Switching, 100 V, 200 mA MMBD914 Multi-Sourced

D3 Diode, Switching, 100 V, 200 mA MMBD914 Multi-Sourced

D4 Diode, Switching, 100 V, 200 mA MMBD914 Multi-Sourced

D5 Diode, Schottky, 70 V, 15 mA 1N5711 ST Microelectronics

D6 Diode, Zener, 0.5-W, 12 V MMSZ5242BT1 ON Semiconductor

J1 Term Block, Eurostyle, 3-Way 39390-0103 Molex

J2 Terminal Block, 2-Way ED120/2DS On-Shore Technology

J3 Header, 0.025 Sq., 2-Way 68001-203HLF FCI BergStik

J4 Terminal Block, 4-Way ED120/4DS On Shore Technology

Q1 N-MOSFET, TO-220, 600 V, 7 A STF9NK60ZD ST Microelectronics

Q2 N-MOSFET, TO-220, 600 V, 7 A STF9NK60ZD ST Microelectronics

Q3 N-MOSFET, LF-PACK, 60 V, 50 A BSC067N06LS3 G Infineon

Q4 N-MOSFET, LF-PACK, 60 V, 50 A BSC067N06LS3 G Infineon

Q5 TRANSISTOR, NPN, 40 V, 200 mA MMBT3904 Multi-Sourced

Q6 JFET, P-CHL, 30 V, 50 mA MMBFJ176 Fairchild

R1 15R0 1% Resistor, SMD, 0805 Generic Multi-Sourced

R2 15R0 1% Resistor, SMD, 0805 Generic Multi-Sourced

R3 N/A Not Fitted

R4 N/A Not Fitted

R5 49k9 1% Resistor, SMD, 0603 Generic Multi-Sourced

R6 49k9 1% Resistor, SMD, 0603 Generic Multi-Sourced

R7 274k 1% Resistor, SMD, 0603 RK73H1JTTD2743F KOA

R8 274k 1% Resistor, SMD, 0603 RK73H1JTTD2743F KOA

R9 49R9 1% Resistor, SMD, 0805 RK73H2ATTD49R9F KOA

R10 49R9 1% Resistor, SMD, 0805 RK73H2ATTD49R9F KOA

R11 49R9 1% Resistor, SMD, 0603 Generic Multi-Sourced

R12 49R9 1% Resistor, SMD, 0603 Generic Multi-Sourced

R13 2R49 1% Resistor, SMD, 0603 Generic Multi-Sourced

R14 2R49 1% Resistor, SMD, 0603 Generic Multi-Sourced

R15 31k6 1% Resistor, SMD, 0603 Generic Multi-Sourced

R16 60k4 1% Resistor, SMD, 0603 Generic Multi-Sourced

R17 10k 1% Resistor, SMD, 0603 Generic Multi-Sourced

R18 10k 1% Resistor, SMD, 0603 Generic Multi-Sourced

R19 49R9 1% Resistor, SMD, 0805 RK73H2ATTD49R9F KOA

R20 10k 1% Resistor, SMD, 0603 Generic Multi-Sourced

R21 26k1 1% Resistor, SMD, 0603 RK73H1JTTD2612F

R22 3k65 1% Resistor, SMD, 0603 Generic Multi-Sourced

R23 348R 1% Resistor, SMD, 0603 Generic Multi-Sourced

R24 6k49 1% Resistor, SMD, 0603 Generic Multi-Sourced

SLUU467–DECEMBER 2010

Table 3 Component Changes for Transformer Synchronization Configuration

Ref. Value Tol. Description Part Number Manufacturer C2 N/A Not Fitted

C8 0.1 μF 10% Capacitor, MLC, 0805, 25 V Generic Multi-Sourced

C13 1000 pF 10% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced

C14 1000 pF 10% Capacitor, MLC, 0603, 50 V Generic Multi-Sourced C15 N/A Not Fitted

C16 N/A Not Fitted

R3 10R0 1% Resistor, SMD, 0603 Generic Multi-Sourced

R4 10R0 1% Resistor, SMD, 0603 Generic Multi-Sourced R31 0R0 Link, SMD, 0805 Generic Multi-Sourced

R32 1k 1% Resistor, SMD, 0603 Generic Multi-Sourced

R33 1k 1% Resistor, SMD, 0603 Generic Multi-Sourced

T3 Pulse Transformer (2:1:1), SMD PA2008NL Pulse

Table 4 Component Changes for Capacitor Synchronization Configuration

Ref. Value Tol. Description Part Number Manufacturer C2 1,000 pF 20% Capacitor, Ceramic T/H, 500 VAC VY1102M35Y5UQ63V0 Vishay-BCcomponents

C8 0R0 Link, SMD, 0805 Generic Multi-Sourced

C13 0R0 Link, SMD, 0603 Generic Multi-Sourced

C14 0R0 Link, SMD, 0603 Generic Multi-Sourced C15 47 pF 10% Capacitor, Ceramic T/H, 500 VAC VY1470K31Y5SQ63V0 Vishay-BCcomponents

C16 47 pF 10% Capacitor, Ceramic T/H, 500 VAC VY1470K31Y5SQ63V0 Vishay-BCcomponents

R3 475R 1% Resistor, SMD, 0603 Generic Multi-Sourced

R4 475R 1% Resistor, SMD, 0603 Generic Multi-Sourced R31 0R0 Link, SMD, 0805 Generic Multi-Sourced

R32 N/A Not Fitted

R33 N/A Not Fitted

T3 N/A Not Fitted

SLUU467–DECEMBER 2010

7 PCB DESIGN

7.1 General

Figure 6 and Figure 7 show the component placement and copper routing of the printed circuit board (PCB) used to

test and characterize this design. The PCB design is based on a double-sided 2-oz. copper-foil layout. A double-

sided PCB provides the flexibility to optimize the layout of the SR FETs and control circuitry. A combination of both

through-hole (T/H) and surface mount devices (SMD) were placed on the top side, and SMD parts on the foil side.

7.2 Grounding

High-frequency transient current associated with switched-power converters induce differential noise voltages in the

ground system. Any voltage differential in the ground can cause spurious operation of the control circuits. The

design dedicates large areas of copper to power ground (PWRGND) and analog ground (AGND). PWRGND

provides the return path for the power circuitry on the primary side of the converter. AGND is used by the control

circuits as the primary-side zero-volt reference. To prevent the AGND from being used as current path by high-

frequency power signals, AGND is referenced to PWRGND at just one location. This single-point grounding

technique forces the high-frequency ground currents generated by the power circuitry to be directed around (as

opposed to through) the quiet ground area of the sensitive control circuits.

It is important to keep the ground connections contiguous. Ground connections should take priority over the routing

of other signals. Where necessary, use vias and T/H components to pass signals to other areas of the board.

These techniques minimize the impedance to high-frequency ground currents to ensure low-noise and reliable

operation.

7.3 Creepage and Clearance

Both the component placement and spacing between the copper areas or the PCB were designed to comply with

the creepage and clearance requirements defined in the UL 60950 safety standard. The board was designed to

meet the requirements for functional isolation for the high-voltage nodes on the primary (ac-line) side of the power

supply, and reinforced isolation between all primary and secondary circuits. To comply with the UL standard 4 mm

of separation was used for the primary-side high-voltage traces, and 8 mm between all primary and secondary side

traces. The default trace separation for low-voltage nodes, as used for control circuits, was 0.3 mm.

7.4 Thermal Considerations

Although this circuit and topology operates with high efficiency, some components on the PCB assembly dissipate

a moderate amount of heat. This may require thermal management initiatives to be implemented when the

converter operates in an enclosure. A thermally conductive path to the surfaces of the enclosure may be required

for these components. Table 5 provides a list of the components that have significant heat dissipation.

Table 5 Parts with High Thermal Dissipation

Reference

Designator Description Dissipation

(1)

Q3, Q4 Secondary SR FETs 1 W

T1 LLC converter transformer 3 W

(1) Estimated dissipation at maximum load and 390 V input voltage.

SLUU467–DECEMBER 2010

Figure 7 PCB Layout; Foil-Side View

SLUU467–DECEMBER 2010

7.6 Power Inputs

7.6.1 HVDC Input

The power input to the circuit requires a high-voltage DC source, capable of supplying up to 420 VDC at 0.5 A max.

This can be provided by a high-voltage adjustable bench supply, the output from a boost PFC circuit, or the rectified

output from a 0–280 V, 1-kVA rated power rheostat. Ensure that the source is either current limited or is fitted with a

fuse with a rating of no more than 1 A.

The HVDC power can be safely applied without the bias supply being present.

7.6.2 Bias Supply

A bias supply of 12–16 VDC at 50 mA is required to operate the converter. The bias power is applied to the J

terminal block and can be supplied from a small adjustable bench supply or 12-V lamp battery. The current

consumption is typically 20 mA when the converter is operating. The polarity is correct when the negative

connection is closest to the J3 connector.

7.7 J3 Header

The PCB design includes a 0.1-in spaced pin header, J3. This header provides on/off control of the converter when

both HVDC input and bias power is applied. This interface can be used to hold down the SS pin (pin 4) until there is

sufficient output voltage for the converter to produce a regulated output. The converter runs only when the open-

circuit voltage at pin 1 of J3 is pulled to GND (J3/Pin 2) using either a jumper or NPN bipolar transistor.

Place a standard 2-pin shorting jumper at J3 to enable the converter to run and produce a DC output. When the

converter is enabled the UCC25600 resonant controller initiates a soft start. Removal of the J3 jumper promptly

shuts down the converter.

The application and removal of the bias supply has the same effect as the removal and replacement of the shorting

jumper at J3.

7.8 Monitoring Primary Current

By making a small modification to the PCB the primary current can be monitored using a current probe. This is also

the current flowing between the series-resonant components, L R and C R.

The PCB layout in Figure 6 shows two test points labeled “SWNODE” located close to and just above the primary

switch, Q2. They are identified TP3 and TP4 on the schematic. The 0.05-in trace connecting these two test points

carries the transformer primary current. This trace can easily be cut, and replaced with a loop of wire connecting

TP3 and TP4. The primary current can then be measured using a clip-on current probe.

SLUU467–DECEMBER 2010

8.3 Typical Output AC Ripple Voltage

0.

0.

0.

0.

0.

0.

1.0 1.5 2.0 2.5 3.0 3.5 4.0 4. Output Current (A)

Output Ripple (V

PP )^330 360 390 420

V (^) IN (V)

Figure 12 Output AC-Ripple with UCC Rectifier Controllers and FETs (2)

0.

0.

0.

0.

0.

0.

1.0 1.5 2.0 2.5 3.0 3.5 4.0 4. Output Current (A)

Output Ripple (V

PP )

330 360 390 420

V (^) IN (V)

Figure 13 Output AC-Ripple with MBR Schottky Rectifiers (2),^ (3)

(2) Peak-to-peak output ripple derived from V (^) OUT rms measurement using an oscilloscope with 20-MHz bandwidth limit. (3) The Schottky diode rectifiers show a higher magnitude of ac rippled at high input voltage (420 V). This is due to an increase in high- frequency ringing from these devices at this input voltage.

9 CIRCUIT WAVEFORMS

The oscilloscope waveforms in this section, captured in Figure 14 through Figure 29, can be identified using

the same color for source identification. The color key to the signals is provided in Table 6.

Table 6 Color Key for Oscilloscope Waveforms

Trace

Color Signal Description

Black Main transformer secondary winding output voltage

Blue Main transformer primary current

Green Upper SR GATE signal waveform (U4/pin 5)

Brown Upper SR SYNC signal waveform (U4/pin 1)

Red Lower SR GATE signal waveform (U5/pin 5)

Orange Lower SR SYNC signal waveform (U5/pin 1)

SLUU467–DECEMBER 2010

9.1 SR Synchronization with Pulse Transformer

Figure 14 Upper and Lower SR Gate Signals at 320-VDC and 1-A Load (f (^) SW <f (^) R )

Figure 15 Upper and Lower Gate Signals at 400-VDC Input and 4.6-A Load (f (^) SWfR )

Figure 16 Upper and Lower SR Gate Signals at 420-VDC Input and 4.6-A Load (f (^) SW >fR )

Figure 17 Upper and Lower SR Gate Signals at 420-VDC Input and 1-A Load (fSW >fR )

Figure 18 Upper/Lower SR Gate and Lower SYNC Signals at 340-VDC Input and 1-A Load (1)/

Figure 19 Upper SR Gate and SYNC Signals at 420-VDC and 3-A Load (2)/

(1) Below resonance the lower SR controller has set the gate signal (Red) low before the negative SYNC pulse (Orange). (2) Above resonance the upper SR controller sets the gate signal (Green) low about the same time as the SYNC pulse (Brown). This forces the gate output low irrespective of whether the UCC24610 controller has sensed the FET is no longer conducting.